Motor controller

ABSTRACT

In order to propose an inexpensive and highly precise motor controller, it is structured so as to detect the position of the rotor on the basis of the difference between the real current differential vector and the reference current differential vector, thereby control the motor without using a rotation position sensor.

BACKGROUND OF THE INVENTION

[0001] The present invention relates to a motor controller.

[0002] To control the speed and torque of a synchronous motor, it isnecessary to detect or infer the pole position. By executing currentcontrol or voltage control on the basis of the detected pole position,the speed and torque of a synchronous motor can be controlled.

[0003] In recent years, a pole position sensorless control system forcontrolling a synchronous motor without detecting the pole position ofthe synchronous motor by a position sensor is proposed.

[0004] For example, the first control method described in JapaneseApplication Patent Laid-Open Publication No. Hei 07-245981 and ElectricSociety, Industry Application Department, National Convention No. 170 inthe 8th years of Heisei is a method for applying an alternating voltageand inferring the pole position on the basis of the parallel componentand orthogonal component (current component in the rotatory coordinatesystem) of the motor current for the voltage and the position of themagnetic pole can be detected without using a pole position sensorduring stopping or at a low speed.

[0005] Further, the second method for superimposing an additionalvoltage described in Japanese Application Patent Laid-Open PublicationNo. Hei 11-150983 and Japanese Application Patent Laid-Open PublicationNo. Hei 11-69884 is a method for realizing no-use of a pole positionsensor within the range from low load to high load during stopping or ata low speed by adding an applied voltage so as to prevent magneticsaturation even in the high torque region.

[0006] Further, the third control method described in JapaneseApplication Patent Laid-Open Publication No. Hei 08-205578 is a methodfor detecting the saliency of a synchronous motor from the mutualrelation between the vector of a voltage applied to the synchronousmotor by the pulse width control (PWM control) and the ripple component(current difference vector) of the motor current for it. The thirdmethod uses a general PWM signal for controlling the voltage of thesynchronous motor, so that there is an advantage that there is no needto load an additional signal for detection.

[0007] Further, the voltage vector means a voltage having the magnitudeand direction decided from a three-phase voltage or d-axis and q-axisvoltages. The same may be said with the current vector and hereinafter,each phase voltage as an element or the d-axis and q-axis voltages andthe voltage vector as a sum total will be explained appropriately.Further, for the synchronous motor, the pole position of the rotor is tobe detected, so that the pole position will be explained hereunder. Fora reluctance motor, the specific position of a rotor having saliency isdetected.

[0008] Further, a control method for detecting the pole position of arotor in the same way as with the aforementioned method on the basis ofthe difference in inductance between the q axis and the q axis using themagnetic saturation characteristic of an induction motor is proposed.

[0009] Therefore, when the aforementioned is to be described together,the pole position and the specific position of the reluctance motor willbe referred to as a rotor position.

SUMMARY OF THE INVENTION

[0010] In the first control method mentioned above, to detect the poleposition by driving the motor, it is necessary to extract a currenthaving the same frequency component as that of the detection alternatingvoltage by a band pass filter such as a notch filter and Fourierintegration. Particularly, when the number of revolutions of the motoris increased, the separation between the input frequency of the motorand the frequency of the detection alternating voltage is difficult anda problem arises that stable driving control at high speed rotation isdifficult. Further, it is necessary to consider so as to prevent effectby the switching characteristic of the invertor. Namely, thecarrier-frequency of the PWM signal is several kHz to 20 kHz, while thefrequency of the detection alternating voltage is low such as severalhundreds Hz, so that during driving control for the motor, noise ofseveral hundreds Hz may be generated.

[0011] Further, the second control method mentioned above is intended toimprove the characteristics for drive-controlling the motor in a stopstate or a low-speed rotation state, and the relation between thecurrent detection timing which is important for drive-controlling themotor at high-speed rotation and the PWM signal is not taken intoaccount, and highly accurate position detection is not taken intoaccount.

[0012] Further, the third control method mentioned above requires, torealize it, to detect the mutual relation between the condition of themotor current and the applied voltage every changing of the PWM signal.Namely, for one period of the carrier, it is necessary to detect themotor current condition at least 6 times and confirm the applied voltagecondition, so that a problem arises that a highly precise controllermust be used.

[0013] An object of the present invention is to propose an inexpensiveand highly precise motor controller.

[0014] Another object of the present invention is to propose a motorcontroller for controlling an AC motor with high precision bysuppressing an increase in motor loss within the wide range from stopstate to high-speed rotation state using one current detector.

[0015] Still another object of the present invention is to propose amotor controller for detecting the rotor position of an AC motor withoutapplying a detection voltage to the AC motor.

[0016] The present invention has an AC motor, a power converter forapplying a voltage to the AC motor by a PWM signal generated bycomparing a command value with a carrier, and a controller for detectingthe rotor position of the AC motor and controlling the command value andis characterized in that the position of the rotor is detected on thebasis of the difference between the real current differential vector andthe reference current differential vector.

BRIEF DESCRIPTION OF THE DRAWINGS

[0017]FIG. 1 is a block diagram of a motor control system of the firstembodiment of the present invention.

[0018]FIG. 2 is a time chart showing the relation between the voltage ofeach phase, PWM signal, and carrier synchronizing signal when thedetection voltage in the first embodiment shown in FIG. 1 is applied.

[0019]FIG. 3 is a low chart of the process executed by the detectioncalculation unit in the first embodiment shown in FIG. 1.

[0020]FIG. 4 is a block diagram showing the relation between input andoutput of the reference current differential calculation unit in thefirst embodiment shown in FIG. 1.

[0021]FIG. 5 is a vector diagram showing the condition of the detectionvoltage vector and current differential vector in the first embodimentshown in FIG. 1.

[0022]FIG. 6 is a vector diagram showing the condition of the detectionvoltage vector and current differential vector in the first embodimentshown in FIG. 1.

[0023]FIG. 7 is a vector diagram showing the condition of the detectionvoltage vector and current differential vector in the first embodimentshown in FIG. 1.

[0024]FIG. 8 is a block diagram of a motor control system showing thesecond embodiment of the present invention.

[0025]FIG. 9 is a flow chart of the process executed by the detectioncalculation unit in the second embodiment shown in FIG. 8.

[0026]FIG. 10 is a flowchart of the current sensor error detectionprocess executed by the current sensor error detection unit in thesecond embodiment shown in FIG. 8.

[0027]FIG. 11 is a block diagram of a motor control system showing thethird embodiment of the present invention.

[0028]FIG. 12 is a block diagram of a motor control system showing thefourth embodiment of the present invention.

[0029]FIG. 13 is a time chart showing the relation between the voltageof each phase, PWM signal, and carrier synchronizing signal in thefourth embodiment shown in FIG. 12.

[0030]FIG. 14 is a time chart and Lissajous waveform diagram showing therelation between the sine wave voltage of each phase and the voltagedifference vector in the fourth embodiment shown in FIG. 12.

[0031]FIG. 15 is a function block diagram of the voltage setting unit inthe fourth embodiment shown in FIG. 12.

[0032]FIG. 16 is a flow chart of the process executed by the h-axiscurrent differential calculation unit in the fourth embodiment shown inFIG. 12.

[0033]FIG. 17 is a vector diagram showing the relation between thecontrol voltage vector, voltage difference vector, and currentdifferential vector in the fourth embodiment shown in FIG. 12.

[0034]FIG. 18 is a time chart and Lissajous waveform diagram showing therelation between the sine wave voltage of each phase and the voltagedifference vector in the fourth embodiment shown in FIG. 12.

[0035]FIG. 19 is a vector diagram showing the relation between thecontrol voltage vector, voltage difference vector, and currentdifferential vector in the fourth embodiment shown in FIG. 12.

[0036]FIG. 20 is a block diagram of a motor control system showing thefifth embodiment of the present invention.

[0037]FIG. 21 is a flow chart of the process executed by the modedecision unit in the fifth embodiment shown in FIG. 20.

[0038]FIG. 22 is a function block diagram of the voltage setting unit inthe fifth embodiment shown in FIG. 20.

[0039]FIG. 23 is a time chart and Lissajous waveform diagram showing therelation between the sine wave voltage of each phase and the voltagedifference vector in the fifth embodiment shown in FIG. 20.

[0040]FIG. 24 is a time chart and Lissajous waveform diagram showing therelation between the sine wave voltage of each phase and the voltagedifference vector in the fifth embodiment shown in FIG. 20.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0041] The first embodiment of the present invention will be explainedwith reference to FIG. 1. The embodiment is structured so as to controla synchronous motor having so-called reverse saliency that the d-axisinductance Ld is smaller than the q-axis inductance Lq without using aposition sensor.

[0042]FIG. 1 is a block diagram of a motor control system for driving asynchronous motor 1 which is the first embodiment by the DC energy of abattery 2.

[0043] The DC voltage of the battery 2 is converted to a three-phase ACvoltage by an inverter 3 which is a power converter and applied to thesynchronous motor 1 which is an AC motor. The applied voltage is decidedby performing the following calculation by a controller 4 which is acontroller composed of a microcomputer.

[0044] The controller 4 calculates the difference of the speed commandvalue ωr input from a speed command generation unit 6 from the detectedmotor speed ω and performs speed control calculations at a speedcontroller 7 on the basis of the difference. The speed controller 7outputs control voltages Vuc, Vvc, and Vwc of each phase on the basis ofthe speed control calculation results. The controller 4 adds detectionvoltages Vus, Vvs, and Vws of each phase, which will be described later,respectively to these control voltages Vuc, Vvc, and Vwc, generatesvoltage command values Vur, Vvr, and Vwr of each phase, and inputs themto a PWM signal generation unit 8.

[0045] The PWM signal generation unit 8 generates PWM signals Pu, Pv,and Pw of each phase corresponding to the voltage command values Vur,Vvr, and Vwr of each phase and supplies them to the inverter 3 and theinverter 3 generates output voltages corresponding to the PWM signalsPu, Pv, and Pw and applies them to the synchronous motor 1.

[0046]FIG. 2 shows the relation between the voltage command values Vur,Vvr, and Vwr of each phase and the PWM signals Pu, Pv, and Pw. The PWMsignal generation unit 8 compares the carrier of triangular waveform andthe voltage command values Vur, Vvr, and Vwr, thereby generates the PWMsignals Pu, Pv, and Pw.

[0047] The PWM signal generation unit 8 internally fetches and sets thevoltage command values Vur, Vvr, and Vwr of each phase at the point oftime (times t1, t3, t5, . . . ) when the carrier takes the maximumvalue, compares them with the carrier, and generates the PWM signals Pu,Pv, and Pw.

[0048] The waveform when the voltage command values Vur, Vvr and Vwr(=the control voltages Vuc, Vvc, and Vwc) to which the detectionvoltages Vus, Vvs, and Vws are not added are fetched is as shown in FIG.2(a).

[0049] On the other hand, when the detection voltages Vus, Vvs, and Vwsare added (applied to the synchronous motor 1), the PWM signalgeneration unit 8 sets the positive and negative detection voltages Vus,Vvs, and Vws every a half period (times t1, t2, t3, . . . ) of thecarrier so as to obtain the waveform shown in FIG. 2(b). Namely, in thesection 1 shown in FIG. 2(b), the PWM signal generation unit 8 adds(applies) the detection voltages Vus, Vvs, and Vws so as to set thedetection voltage vector in the detection voltage direction θv whichwill be described later. Further, in the section 2 shown in FIG. 2(b),the PWM signal generation unit 8 adds the detection voltages Vus, Vvs,and Vws so as to apply the detection voltage vector in the oppositedirection (in the direction different by 180 degrees) to the detectionvoltage direction.

[0050]FIG. 3 is a flow chart showing the process to be executed by adetection voltage calculation unit 10 to realize it.

[0051] Step 101

[0052] The detection voltage direction θv is obtained by calculation of2θc/2. The reason for the voltage direction will be described later byreferring to FIGS. 5 to 7.

[0053] Step 102

[0054] The voltage application timing is judged and the process isbranched. Namely, as shown in FIG. 2, when the times t1, t3, . . . arejudged as a point of time when the carrier reaches the maximum value,the process is branched to Step 103 and when the times t2, t4, . . . arejudged as a point of time when the carrier reaches the minimum value,the process is branched to Step 104.

[0055] Step 103

[0056] To set the vector Vs of detection voltage applied to thesynchronous motor 1 in the section 1 in the detection voltage directionθv (positive direction), the detection voltages Vus, Vvs, and Vws ofeach phase are calculated.

[0057] Step 104

[0058] To set the vector Vs of detection voltage applied to thesynchronous motor 1 in the section 2 in the detection voltage directionθv (negative direction, that is, direction of θv+π), the detectionvoltages Vus, Vvs, and Vws of each phase are calculated.

[0059] In FIG. 3, Vs0 for deciding the magnitude of the detectionvoltage vector Vs is set to ½, thus the voltage of each phase isdecided. The reason is that the voltage difference between the sections1 and 2 is defined a the real detection voltage vector Vs. Further, itis desirable to set Vs0 to a small value as far as possible as long asthe variation of current can be detected. Further, in this case, on thebasis of the α axis of the α-β static coordinate system having theorthogonal α axis and s axis, the phase or direction is decided and theU phase is set on the α axis. Therefore, the directions of the V and Wphases are directions of 2π/3 and 4π/3 to the α axis respectively.

[0060] Next, the detection method for the rotor position in the firstembodiment shown in FIG. 1 will be explained.

[0061] As a current sensor 5 u for detecting the U-phase current of thesynchronous motor 1, an inexpensive current transformer CT for detectingonly the AC component of a current flowing in the U phase is used. Bydoing this, only the pulsating component of a current by the PWM signalis detected.

[0062] A current detection unit 9 of the controller 4 fetches anddetects a U-phase current iu output from the current transformer CT inthe timing coinciding with a carrier synchronous signal P1 synchronizedwith the maximum value and minimum value of the carrier.

[0063] A current differential calculation unit 11 obtains the variationof the U-phase current iu from the detection voltage vector, that is,the U-phase current difference Δiu as shown below. The currentdifferential calculation unit 11 calculates the current difference Δiu1in the section 1 shown in FIG. 2 from the difference between the U-phasecurrent iu1 fetched by the current detection unit 9 at the point of timeof the maximum value of the carrier (for example, time t1) and theU-phase current iu2 fetched at the point of time of the next minimumvalue of the carrier (time t2). Further, the current differentialcalculation unit 11 calculates the current difference Δiu2 in thesection 2 shown in FIG. 2 from the difference between the U-phasecurrent iu2 and the U-phase current iu3 fetched at the point of time ofthe next maximum value of the carrier (time t3). The current differencesΔiu1 and Δiu2 are affected by the control voltages Vuc, Vvc, and Vwc,the detection voltage vector, and the counter electromotive force of thesynchronous motor 1. However, in consideration of the difference betweenthe current differences Δiu1 and Δiu2, when the applied voltage andcounter electromotive force are the same, their effects are canceled.

[0064] Therefore, as explained by referring to FIG. 2, in the sections 1and 2, when the control voltages Vuc, Vvc, and Vwc are applied in thesame value and only the detection voltage vectors Vs are applied indifferent values, the U-phase current difference Δiu which is thedifference between the current differences Δiu1 and Δiu2 is affectedonly by the difference in the detection voltage vector Vs between thesections 1 and 2. Namely, the variation of the U-phase current iu to thedetection voltage vector Vs and the U-phase current difference Δiu canbe detected quite independently of the control voltages Vuc, Vvc, andVwc. Hereinafter, the difference in the detection voltage vector betweenthe sections 1 and 2 is called a detection voltage vector Vs.

[0065] Meanwhile, when the rotor position θ and the d-axis and q-axisinductances of the synchronous motor 1 are known, the variation of theU-phase current iu to the detection voltage vector Vs can be obtained bycalculation. This value is assumed as a U-phase reference currentdifference Δicu. Actually, instead of the rotor position θ, the inferredrotor position θc calculated by the controller 4 is known, so thatassuming that the inferred rotor position θc agrees with the rotorposition θ, the U-phase reference current difference Δicu is obtained bya reference current differential calculation unit 12. In this process,as shown in FIG. 4, when a table is prepared for the inferred rotorposition θc, the U-phase reference current difference Δicu can beobtained simply. This obtaining method will be described later togetherwith the vector diagrams of FIGS. 5 to 7.

[0066] The difference between the detected U-phase current differenceΔiu and the U-phase reference current difference Δicu (hereinafter, thisis called a U-phase detection current difference Δisu) indicates avariation (difference) between the inferred rotor position θc and therotor position θ, so that a position detection unit 13 controlsconverging by using a control means such as proportion-integrationcalculations so as to set the difference to 0.

[0067] It is an important point of the present invention to make therelation thereof clear and it will be described later by referring tothe vector diagrams of FIGS. 5 to 7.

[0068] The inferred rotor position θc obtained as mentioned above isinput to a speed detection unit 14 and used to obtain the motor speed ωfrom the variation thereof. Further,the inferred rotor position θc isinput to the speed control unit 7 and also used to output the controlvoltage vector obtained by the speed control unit 7 to the controlvoltages Vuc, Vvc, and Vwc of each phase by coordinate conversion.

[0069] Next, the detection of the rotor position θ in the motor controlsystem shown in FIG. 1 will be explained by referring to FIG. 5.

[0070]FIG. 5 shows a state that the d-q axis rotatory coordinate systemthat the pole position is on the d axis rotates from the α axis by therotor position θ and the inferred rotor position θc of the controller 4is larger than the real rotor position θ and different from the realrotor position θ. The ellipse indicated by a solid line that the d axisis a long axis and the q axis is a short axis indicates a Lissajouswaveform of the current differential vector Δi to the detection voltagevector Vs when the detection voltage vector Vs makes one revolution from0 to 2π. Therefore, when the inductances of the d axis and q axis of thesynchronous motor 1 are the set values, the actually-generated currentdifferential vector Δi to the detection voltage vector Vs moves on thesolid-line ellipse. On the other hand, when the inductances of the daxis and q axis of the synchronous motor 1 are as set and the rotorposition of the synchronous motor 1 coincides with the inferred rotorposition θc inferred by the controller 4, the current differentialvector Δi moves on the ellipse indicated by a dashed line that the dcaxis is a long axis and the qc axis is a short axis. This is called areference current differential vector Δic.

[0071] In this state, as shown in FIG. 5, the change condition ofcurrent when the detection voltage vector Vs is applied in the detectionvoltage direction θv, that is, in the direction of 2θc+π/2 will beexplained.

[0072] When the detection voltage vector Vs is applied in the phase θvdirection, the current differential vector Δi actually generated, asshown in FIG. 5, is a vector on the Lissajous waveform line indicated bya solid line and expressed by the following formula.

Δi=Δiα+jΔiβ  (Formula 1)

[0073] where j means an imaginary axis and Δiα and Δis mean thefollowing formulas.

Δiα=Δids·cos θΔiqs·sin θ  (Formula 2)

Δis=Δids·sin θ+Δiqs·cos θ  (Formula 3)

Δids=Vs 0·cos(θv−θ)Δt/Ld  (Formula 4)

Δiqs=Vs 0·sin(θv−θ)Δt/Ld  (Formula 5)

[0074] where Ld and Lq indicate the inductances of the d axis and q axisof the synchronous motor 1 respectively and Vs0 indicates the magnitudeof the detection voltage (length of the detection voltage vector Vsshown in FIG. 5). Therefore, Formula 2 and Formula 3 are expressed asfollows. $\begin{matrix}{{\Delta \quad i\quad \alpha} = \quad {{\left( {1/2} \right) \cdot V}\quad s\quad {0 \cdot \Delta}\quad t\left\{ {{\left( {{{1/L}\quad d} + {{1/L}\quad q}} \right)\cos \quad \theta \quad v} + {\left( {{{1/L}\quad d} - {{1/L}\quad q}} \right){\cos \left( {{\theta \quad v} - {2\theta}} \right)}}} \right\}}} & \left( {{Formula}\quad 6} \right) \\{{\Delta \quad i\quad s} = \quad {{\left( {1/2} \right) \cdot V}\quad s\quad {0 \cdot \Delta}\quad t\left\{ {{\left( {{{1/L}\quad d} + {{1/L}\quad q}} \right)\sin \quad \theta \quad v} - {\left( {{{1/L}\quad d} - {{1/L}\quad q}} \right){\sin \left( {{\theta \quad v} - {2\theta}} \right)}}} \right\}}} & \left( {{Formula}\quad 7} \right)\end{matrix}$

[0075] In the same way, when the real rotor position coincides with theinferred rotor position θc inferred by the controller 4, the referencecurrent differential vector Δic generated by applying the detectionvoltage vector Vs in the phase θv direction is on the Lissajous waveformindicated by a dashed line. In FIG. 5, the detection voltage vector Vsis closer to the qc axis than the q axis, so that the reference currentdifferential vector Δic is a vector closer to the detection voltagevector Vs than the current differential vector Δi and expressed by thefollowing formula.

Δic=Δiαc+jΔiβc  (Formula 8)

[0076] where Δiαc and Δiβc mean the following formulas.

Δiαc=Δidsc·cos θc−Δiqsc·sin θc  (Formula 9)

Δiβc=Δidsc·sin θc+Δiqsc·cos θc  (Formula 10)

Δidsc=Vs0·cos(θv−θc)Δt/Ldc  (Formula 11)

Δiqsc=Vs0·sin(θv−θc)Δt/Ldc  (Formula 12)

[0077] where Ldc and Lqc indicate the inductances of the reference daxis and q axis of the synchronous motor 1 set by the controller 4.Therefore, Formula 9 and Formula 10 are expressed as follows.$\begin{matrix}{{\Delta \quad i\quad \alpha \quad c} = \quad {{\left( {1/2} \right) \cdot V}\quad s\quad {0 \cdot \Delta}\quad t\left\{ {{\left( {{{1/L}\quad d\quad c} + {{1/L}\quad q\quad c}} \right)\cos \quad \theta \quad v} + {\left( {{{1/L}\quad d\quad c} - {{1/L}\quad q\quad c}} \right){\cos \left( {{\theta \quad v} - {2\theta \quad c}} \right)}}} \right\}}} & \left( {{Formula}\quad 13} \right) \\{{\Delta \quad i\quad s\quad c} = \quad {{\left( {1/2} \right) \cdot V}\quad s\quad {0 \cdot \Delta}\quad t\left\{ {{\left( {{{1/L}\quad d\quad c} + {{1/L}\quad q\quad c}} \right)\sin \quad \theta \quad v} - {\left( {{{1/L}\quad d\quad c} - {{1/L}\quad q\quad c}} \right){\sin \left( {{\theta \quad v} - {2\theta \quad c}} \right)}}} \right\}}} & \left( {{Formula}\quad 14} \right)\end{matrix}$

[0078] In this case, the U-phase reference current difference Δicu shownin FIG. 4 is the U-phase contribution of Δiαc and a value proportionalto Δiαc, so that a table can be prepared by calculation including thedetection voltage direction θv. Further, the detection voltage directionθv is set to a value of (2θc+π/2), though it will be described later.Therefore, Formula 3 is expressed as follows, so that a table forobtaining the U-phase reference current difference Δicu is prepared onthe basis of Formula 15. $\begin{matrix}{{\Delta \quad i\quad \alpha \quad c} = \quad {{\left( {1/2} \right) \cdot V}\quad s\quad {0 \cdot \Delta}\quad {t\left( {{{1/L}\quad d\quad c} + {{1/L}\quad q\quad c}} \right)}{\cos \left( {{2\theta \quad c} + {\pi/2}} \right)}}} & \left( {{Formula}\quad 15} \right)\end{matrix}$

[0079] Next, the difference between the current differential vector Δiand the reference current differential vector Δic and the detectioncurrent differential vector Δis will be examined. Further, theinductances Ldc and Lqc of the reference d axis and q axis of thesynchronous motor 1 are respectively different from the inductances Ldand Lq of the real d axis and q axis and a method taking it into accountmay be used together. However, in this case, Ldc=Ld and Lqc=Lq areassumed to be held.

[0080] The following may be obtained using Formula 1 to Formula 4.

Δis=Δi−Δic≡Δiαs+jΔiss  (Formula 16)

[0081] where Δiαs and Δiss are given the following formulas.$\begin{matrix}{{{\Delta i\alpha}\quad s} = {{\left( {1/2} \right) \cdot V}\quad s\quad {0 \cdot \Delta}\quad {t\left( {{{1/L}\quad d} - {{1/L}\quad q}} \right)}\left\{ {{\cos \left( {{\theta \quad v} - {2\theta}} \right)} - {\cos \left( {{\theta \quad v} - {2\theta \quad c}} \right)}} \right\}}} & \left( {{Formula}\quad 17} \right) \\{{\Delta \quad i\quad s\quad c} = {{\left( {1/2} \right) \cdot V}\quad s\quad {0 \cdot \Delta}\quad {t\left( {{{1/L}\quad d} - {{1/L}\quad q}}\quad \right)}\left\{ {{\sin \left( {{\theta \quad v} - {2\theta}} \right)} - {\sin \left( {{\theta \quad v} - {2\theta \quad c}} \right)}} \right\}}} & \left( {{Formula}\quad 18} \right)\end{matrix}$

[0082] Then, when (2θc+π/2) is substituted for the detection voltagedirection θv, Formula 17 and Formula 18 can be expanded as shown below.Further, (1/2).Vs0.Δt(1/Ld−1/Lq) is assumed as a constant of K0.$\begin{matrix}{\begin{matrix}{{\Delta \quad i\quad \alpha \quad s} = \quad {{K0}\left\{ {{\cos \left( {{\theta \quad v} - {2\theta}} \right)} - {\cos \left( {{\theta \quad v} - {2\theta \quad c}} \right)}} \right\}}} \\{= \quad {{K0}\left\{ {{\cos \left( {{2\theta \quad c} - {2\theta} + {\pi/2}} \right)} - {\cos \left( {\pi/2} \right)}} \right\}}} \\{= \quad {{{- {K0}} \cdot \sin}\quad 2\left( {{\theta \quad c} - \theta} \right)}} \\{= \quad {{{- {K0}} \cdot \sin}\quad 2\left( {\theta - {\theta \quad c}} \right)}}\end{matrix}\quad} & \left( {{Formula}\quad 19} \right) \\{\begin{matrix}{{\Delta \quad i\quad \beta \quad s} = \quad {{- {K0}}\left\{ {{\sin \left( {{\theta \quad v} - {2\theta}} \right)} - {\sin \left( {{\theta \quad v} - {2\theta \quad c}} \right)}} \right\}}} \\{= \quad {{- {K0}}\quad {K0}\left\{ {{\sin \left( {{2\theta \quad c} - {2\theta} + {\pi/2}} \right)} - {\sin \left( {\pi/2} \right)}} \right\}}} \\{= \quad {{K0} \cdot \left\{ {1 - {\cos \quad 2\left( {{\theta \quad c} - \theta} \right)}} \right\}}}\end{matrix}\quad} & \left( {{Formula}\quad 20} \right)\end{matrix}$

[0083] The vector expressed by Formula 19 and Formula 20 is thedetection current differential vector Δis. Therefore, as shown in FIG.5, when the inferred rotor position θc is larger than the real rotorposition θ, if the detection voltage vector Vs is applied assuming thatthe detection voltage direction θv is (2θc+π/2), the detection currentdifferential vector Δis is a vector in the direction close to thenegative direction of the α axis. Particularly, in consideration ofFormula 19, when the detection voltage vector Vs is applied assumingthat the detection voltage direction θv is (2θc+π/2), the α axiscomponent Δiαs of the detection current differential vector Δis is avalue proportional to sin 2(θ−θc). Therefore, when the α axis componentΔiαs is set to 0, the inferred rotor position θc can coincide with thereal rotor position θ. Further, in this embodiment, the U phase of thesynchronous motor 1 coincides with the α axis, the α axis component Δiαsof the detection current differential vector Δis is proportional to theU-phase detection current differential vector Δisu. Therefore, in FIG.5, the U-phase detection current differential vector Δisu is negativeand it means that the inferred rotor position θc is larger than the realrotor position θ, so that by performing control calculations so as toreduce the inferred rotor position θc, the inferred rotor position θccan be brought close to the real rotor position θ. Such calculations areperformed by the position detection unit 13.

[0084]FIG. 6 shows the relation between the detection voltage vector Vsand the current differential vectors Δi, Δic, and Δis for it when theinferred rotor position θc is close to the real rotor position θ in thisway. Since the inferred rotor position θc is made smaller, it is foundthat the direction θv of the detection voltage vector Vs is smaller thanthat shown in FIG. 5. Therefore, the current differential vector Δi andthe reference current differential vector Δic respectively move in thedifferent directions from those shown in FIG. 5, while the detectioncurrent differential vector Δis is directed almost in the negativedirection of the α axis and the magnitude thereof is made smaller.Therefore, it is found that the inferred rotor position θc is close tothe real rotor position θ. On the basis of the U-phase detection currentdifferential vector Δisu proportional to the α axis component thereof,the inferred rotor position θc can coincide with the real rotor positionθ by control calculation.

[0085] When the inferred rotor position θc is small for the real rotorposition θ as shown in FIG. 7, the detection current differential vectorΔis is directed close to the positive direction of the α axis, so thatthe inferred rotor position θc can coincide with the real rotor positionθ by control calculation in the same way. These relations can be derivedfrom Formula 19 and Formula 20.

[0086] In this embodiment, the rotor position can be inferred preciselyusing one inexpensive current sensor, so that compared with aconventional controller using a plurality of current sensors, aninexpensive position sensorless controller can be realized. Moreover,the inference can be executed by a calculation process on the basis of aone-phase current, so that the controller 4 can be realized using aninexpensive microprocessor.

[0087] Further, the current sensor 5 u may detect only the pulsatingcomponent of a current on the basis of the PWM signal, so that a signalhaving a frequency component in the neighborhood of the carrierfrequency is input to the current detection unit 9, thus the resolutionof current detection can be improved. By doing this, the magnitude ofthe detection voltage vector Vs can be reduced and there is an advantagethat the effect by addition of the detection voltage can be reducedgreatly.

[0088]FIG. 8 is a block diagram of a motor control system showing thesecond embodiment of the present invention. In this embodiment, ascompared with the first embodiment, the method for applying thedetection voltage vector Vs is different in a point that two currentsensors 5 v and 5 w are used for current control for a torque commandinstead of a speed command. The method for inferring the rotor positionusing one current sensor 5 u is the same as that of the firstembodiment, so that duplicate explanation will be omitted.

[0089] This embodiment is suitable for motor control of an electric carfor generating torque proportional to a torque command τr according tothe stepping depth of an accelerator pedal.

[0090] When the torque command τr is input to the current command unit16, the current command unit 16 calculates a d-axis current commandvalue idr for controlling the magnetic flux of the synchronous motor 1and a q-axis current command value idq orthogonal to it on the basis ofthe torque command τr and the motor speed ω. The calculation may beperformed so as to obtain from a table prepared by calculating thed-axis current command value idr and the q-axis current command valueidq orthogonal to it beforehand so as to minimize the loss of thedriving system of the synchronous motor 1 for the torque command τr andthe motor speed ω. The d-axis current command value idr and the q-axiscurrent command value idq obtained here are input to the current controlunit 17.

[0091] Further, a V-phase current iv and a W-phase current iw which aredetected by the current sensors 5 v and 5 w are converted from analogueto digital by the current detection unit 15 and fetched inside thecontroller 4 as a digital amount. Thereafter, by the coordinateconversion unit 19, these currents are coordinate-converted from thestatic coordinate system to the d-q axis rotatory coordinate systemrotating in the same way as with the rotor using the inferred rotorposition θc obtained by the position detection unit 13 and a d-axiscurrent id and a q-axis current id are obtained.

[0092] The d-axis current id and the q-axis current id are input to thecurrent control unit 17, and the feedback control calculation by thedifference between the d-axis current command value idr and the d-axiscurrent id is performed by the current control unit 17, thus a d-axiscontrol voltage Vdc is decided, and the feedback control calculation bythe difference between the q-axis current command value iqr and theq-axis current iq is performed, thus a q-axis control voltage Vqc isdecided. The control calculation is generally the proportion-integrationcalculation. Further, as a method for correcting the counterelectromotive force accompanying rotation of the synchronous motor 1,non-interference control according to the motor speed ω may be usedtogether.

[0093] From the viewpoint of control, when the detection voltages Vqsand Vds are ignored, the d-axis voltage command value Vdr (=d-axiscontrol voltage Vdc) and the q-axis voltage command value Vqr (=q-axiscontrol voltage Vqdc) are converted from the d-q axis rotatorycoordinate system to the α-β axis static coordinate system by thecoordinate conversion unit 18 and 3-phase voltage commands Vur, Vvr, andVwr are output. By this addition of the current control system, thed-axis current id can coincide with the d-axis current command value idrand the q-axis current id can coincide with the d-axis current commandvalue idr respectively at a rapid response speed.

[0094] In the second embodiment, the position sensorless control systemcan realize rapid-response torque control.

[0095] Further, the detection voltage calculation unit 20 in the secondembodiment is structured so as to apply the detection voltages Vds andVqs in the d-q axis coordinate system. The processing method executed bythe detection voltage calculation unit 20 will be explained by referringto FIG. 9.

[0096]FIG. 9 is a flow chart of the process executed by the detectionvoltage calculation unit 20.

[0097] Step 111

[0098] The detection voltage direction θv is obtained by calculation of(θc+π/2). The reason is that a detection voltage is applied to the d-qaxis rotatory coordinate system and as shown in the vector diagrams inFIGS. 5 to 7, the phase difference between the α-s axis staticcoordinate system which is a static coordinate system and the dc-qc axisrotatory coordinate system is θc.

[0099] Step 112

[0100] In the same way as with Step 102 mentioned above, the detectionvoltage calculation unit 20 judges the timing of voltage application andbranches the process. In this case, the detection voltage is changedevery a half period of the carrier. However, when the application periodof the control voltage is equal to two periods of the carrier, thejudgment change at Step 112 may be executed every one period of thecarrier.

[0101] Step 113

[0102] The detection voltages Vds and Vqs applied at the point of timewhen the carrier is maximized are calculated.

[0103] Step 114

[0104] The detection voltages Vds and Vqs applied at the point of timewhen the carrier is minimized are calculated.

[0105] The processing method shown in FIG. 9 has an advantage comparedwith the processing method shown in FIG. 3 that there are very fewcalculation contents.

[0106] Further, the second embodiment is added with a current sensorerror detection unit 21. Generally, a method for detecting an error ofthe current sensor using that the sum of three-phase currents is 0 isknown. However, in the second embodiment shown in FIG. 8, in the sameway as with the first embodiment shown in FIG. 1, the current sensor 5 uof U-phase is structured so as to use an inexpensive sensor having afunction for detecting only the AC amount, so that to detect existenceor no-existence of an error of the current sensor, a new detectionmethod must be designed.

[0107]FIG. 10 is a flowchart of the current sensor error detectionprocess executed by the current sensor error detection unit 21.

[0108] Step 121

[0109] The U-phase current difference Δiu is obtained from thedifference between the U-phase current iu(n) at the time t(n) and theU-phase current iu(n−1) at the time t(n−1). In this case, the time (n)means the point of time of the minimum value of the carrier andconcretely, it is equivalent to the times t2 and t4 shown in FIG. 2.Further, the time (n−1) means the point of time of the maximum value ofthe carrier and in the same way, it is equivalent to the times t1 and t3shown in FIG. 2. Further, with respect to the U-phase current, thecurrent sensor 5 u for detecting only the AC component is used, so thata value different from the current actually flowing is detected, whilethe U-phase current difference Δiu which is a fluctuation component isthe same as the real value.

[0110] Step 122

[0111] The U-phase current difference Δiv is obtained in the same way.

[0112] Step 123

[0113] The W-phase current difference Δiw is obtained in the same way.

[0114] With respect to the V phase and W phase, for the purpose ofexecution of current control during stop and at a low speed, a currentsensor capable of also detecting the DC component may be used.

[0115] Step 124

[0116] The sum total Δi0 of 3-phase current differences is calculated.In a general case that a zero-phase current does not flow in thesynchronous motor 1, the sum total of 3-phase currents is 0, so that thesum total Δi0 of 3-phase current differences is also 0.

[0117] Step 125

[0118] Whether the sum total Δi0 of current differences is less than apredetermined decided value Δij or not is decided. When the sum totalΔi0 is less than the decided value Δij, the current sensor errordetection unit 21 judges that the current sensor is normal and ends theprocess, and when the sum total Δi0 is the decided value Δij or more,the process is branched to Step 126.

[0119] Step 126

[0120] The current sensor error detection unit 21 generates a currentsensor error signal Sc and inputs it to the PWM signal generation unit8. The PWM signal generation unit 8, when the current sensor errorsignal Sc is input, stops generation of the PWM signals Pu, Pv, and Pwand stops the synchronous motor 1.

[0121] When the current sensors 5 u, 5 v, and 5W enter an error statelike this, the synchronous motor 1 is stopped, thus the high reliabilityof the position sensorless control system for executing highly efficientcurrent control is ensured.

[0122]FIG. 11 is a block diagram of a motor control system showing thethird embodiment of the present invention. This embodiment is anembodiment that a motor control system having the equivalent performanceto that of the motor control system of the second embodiment isstructured at a low price. For the constitution duplicated with theaforementioned embodiment, the explanation will be omitted.

[0123] The third embodiment has a constitution that one current sensor 5x for detecting the input current of the inverter 3 is used instead ofthe current sensors 5 v and 5 w, and the input current iDC detected bythe current sensor 5 x and the PWM signals Pu, Pv, and Pw of each phaseare input to a phase current separation unit 22, and the V-phase currentiv and the W-phase current iw are obtained by calculation.

[0124] From the logic of the PWM signals Pu, Pv, and Pw of three-phase,the relation between the input current iDC and the currents of eachphase is found. For example, when the PWM signal Pu is on the high leveland the PWM signals Pv and Pw are on the low level, the power element onthe upper side of the V phase and the power elements on the lower sideof the U phase and W phase in the 3-phase bridge circuit of the inverter3 are turned on, so that the input current iDC agrees with the positiveV-phase current. Further, when the PWM signals Pu and Pv are on the highlevel and the PWM signal Pw is on the low level, the power elements onthe upper side of the U phase and V phase and the power element on thelower side of the W phase are turned on, so that the input current iDCagrees with the negative W-phase current. By tabling the relationbetween this pattern of the PWM signals Pu, Pv, and Pw and the phasecurrents, the current of each phase can be obtained on the basis of thedetected input current iDC and the PWM signals Pu, Pv, and Pw. The phasecurrent separation unit 22 obtains the V-phase current iv and theW-phase current iw on the basis of this relation between the inputcurrent iDC, the PWM signals Pu, Pv, and Pw, and each phase current.This constitution can reduce the number of current sensors to be used.Further, the current sensor 5 u for position detection is used to detecta current of one phase (U phase in this case) every predetermined timingsynchronized with the carrier.

[0125] According to the controller of the third embodiment, arapid-response position sensorless control system can be realized at alow price.

[0126]FIG. 12 is a block diagram of a motor control system showing thefourth embodiment of the present invention. This embodiment isstructured so as to detect the phase current without applying adetection voltage at the timing showing the same phenomenon as that whenthe detection voltage is applied, thereby realize detection of the rotorposition using the saliency of the synchronous motor 1, prevent anincrease in the loss due to an increase in noise and pulsation of thecurrent by applying the detection voltage, and realize a rapid-responseposition sensorless control system at a low price.

[0127] Therefore, the current differential vector is detected using twocurrent sensors. The fourth embodiment shown in FIG. 12 constitutes acurrent control system for the torque command τr in the same way as withthe second embodiment shown in FIG. 8. However, a method for applying nodetection voltage, obtaining a voltage vector instead of it, andinferring the rotor position θc is different. For the complicateconstitution with the aforementioned embodiment, the explanation will beomitted.

[0128] In the fourth embodiment, the PWM signal generation unit 8generates a carrier synchronizing signal P2 for current detection attiming different from that of the carrier synchronizing signal P1 of theaforementioned embodiment. The carrier synchronizing signal P2, as shownin the time chart in FIG. 13, is generated so as to detect the currentof each phase at the times ta, tb, tc, and td when the carrier takes anintermediate value.

[0129] The current detection unit 15 fetches the V-phase current iv andthe W-phase current iw at the generation timing of the carriersynchronizing signal P2. The application condition of the voltage ofeach phase at this time will be explained by referring to FIG. 13.Further, during the period from the time t1 to t5, it is assumed thatthe voltages (control voltages Vur, Vvr, and Vwr) to be applied to eachphase are not changed.

[0130] The mean voltage of each phase during the period from the time t1to t5 (or t3) is naturally Vur, Vvr, and Vwr respectively. However, asthe PWM signals Pu, Pv, and Pw, show, in the section A between the timesta and tb, the U-phase voltage and V-phase voltage are maximum valuesand the W-phase voltage is a negative value, while in the section Bbetween the times tb and tc, the U-phase voltage is close to 0, and theV-phase voltage is a negative value, and the W-phase voltage is aminimum value. Namely, between the section A and the section B, there isa difference in the voltage vector to be applied. This is called avoltage difference vector ΔVs. In the time chart of the first embodimentshown in FIG. 2, the positive and negative detection voltage vectors areadditionally applied in the sections 1 and 2 so as to generate adifference in the voltage vector. However, when the current detectiontiming is set as in the fourth embodiment, an equivalent state to thatwhen the detection voltage is applied can be set.

[0131] Next, the relation between the control voltage vector Vc and thevoltage difference vector ΔVs will be explained by referring to FIG. 14.FIG. 14(a) shows waveforms when the voltage of each phase for the phaseof the control voltage vector is a sine waveform. In this case, theLissajous waveforms of the voltage vectors in the sections A and B arerespectively as shown in (b) and (c) and formed like a bulged triangle.In this case, the arrows shown in (b) and (c) indicate voltage vectorswhen the phase of the control voltage vector is 150 degrees. Further,the mean voltage vector in the sections A and B is a circle as shown in(d) and the phase of the mean voltage vector indicated by the arrow is150 degrees. The average of the two voltage vectors indicated by thearrows in (b) and (c) is the mean voltage vector in (d). The Lissajouswaveform of the mean voltage vector is naturally circular because itindicates a voltage vector of a sine wave.

[0132] The Lissajous waveform of the voltage difference vector ΔVs is asshown in (e) and the voltage difference vector ΔVs when the phase of thecontrol voltage vector is 150 degrees is directed in the direction of−120 degrees. When the phase of the control voltage vector is set as atransverse axis, and the absolute value ΔVs0 of the voltage differencevector ΔVs and the phase θv thereof are set as an ordinate axis, and theLissajous waveform of the voltage difference vector ΔVs shown in (e) isconverted, it is as shown in (f). The absolute value ΔVs0 of the voltagedifference vector ΔVs pulsates in a period of {fraction (1/6 )}times ofone period of the phase of the control voltage vector and the phase θvof the voltage difference vector ΔVs rotates two times in one period ofthe phase of the control voltage vector. When the control voltages Vur,Vvr, and Vwr of each phase are decided, the voltage difference vectorΔVs is decided uniquely and the control voltage vector can be calculatedby tabling the phase thereof.

[0133] Therefore, in the fourth embodiment, when the control voltages Vdand Vq of the d and q axes decided by the current control unit 17 andthe inferred rotor position θc output from the position detection unit13 are input to the voltage setting unit 25, the voltage command valuesVur, Vvr, and Vwr of each phase are calculated and the absolute valueΔVs0 of the voltage difference vector ΔVs and the phase θv thereof canbe obtained by the table. These values are output to perform positiondetection calculations.

[0134] The voltage setting unit 25 will be explained by referring to thefunction block diagram shown in FIG. 15. The voltage vector calculationunit 27 obtains the absolute value Vc0 of the control voltage vector andthe vector phase δ from the dc axis on the basis of the control voltagesVd and Vq. By adding the vector phase δ and the inferred rotor positionθc, the phase θvc from the α axis of the control voltage vector isobtained, and then the single-phase voltage table 28 is referred to, andthe single-phase voltages vu, vv, and vw which are the basis for theapplied voltages of each phase are obtained. In this case, the waveformsof sine wave shown in FIG. 14(a) are tabled.

[0135] Further, the absolute value θVs0 of the voltage difference vectorΔVs and the phase θv thereof, as shown in FIG. 14(f), are decided by theabsolute value Vc0 of the control voltage vector and the phase θvcthereof. Then, in the single-phase voltage table 28, the phase θv of thevoltage difference vector for the phase θvc and the unit voltagedifference vs which is the absolute value ΔVs0 of the voltage differencevector when the absolute value Vc0 of the control voltage vector is 1 Vare tabled, thus the unit voltage difference vs and the phase θv arecalculated. The magnitude of voltage is proportional to the absolutevalue Vc0 of the control voltage vector, so that the products of theabsolute value Vc0 of the control voltage vector, the single-phasevoltages vu, vv, and vw, and the unit voltage difference vs arecalculated respectively by the multiplication unit 29 and the voltagecommand values Vur, Vvr, and Vwr of each phase and the absolute valueΔVs0 of the voltage difference vector are obtained.

[0136] Next, a method for inferring the rotor position from the voltagedifference vector will be explained. In the first embodiment shown inFIG. 1, there is a degree of freedom for applying the detection voltagevector Vs in an optional direction and it is used. Namely, the directionof the detection voltage vector Vs is decided so that a detectioncurrent difference Δis for an error of the rotor position appears in aspecific direction (in the first embodiment shown in FIG. 1, thedirection of the α axis). However, the voltage difference vector for thedetection voltage vector Vs is uniquely decided when the control voltagevector is decided, so that it cannot be set in an optional direction.Therefore, the fourth embodiment has a constitution that inversely forthe decided voltage difference vector and the inferred rotor positionθc, the direction that the detection current difference Δis for an errorof the rotor position appears is identified and control for setting thecomponent of the detection current difference Δis in the direction to 0is executed. Further, the direction that the detection currentdifference Δis appears is set as an h axis here and the phase thereof isset to θh.

[0137] When the calculation method for Formula 1 to Formula 20 is used,the phase θh of the h axis is obtained by the following formula.

θh=2θc−θv+π/2  (Formula 21)

[0138] The h-axis phase calculation unit 26 in the fourth embodimentcalculates Formula 21 on the basis of the inferred rotor position θc andthe voltage difference vector phase θv and outputs the phase θv of the haxis. In this case, the h-axis reference current difference Δihc whichis the θh component in the h-axis direction of the reference currentdifferential vector Δic is obtained by the following formula.$\begin{matrix}{\begin{matrix}{{\Delta \quad i\quad c} = \quad {{\Delta \quad i\quad \alpha \quad {c \cdot \cos}\quad \theta \quad h} + {\Delta \quad i\quad \beta \quad {c \cdot \sin}\quad \theta \quad h}}} \\{= \quad {{\left( {1/2} \right) \cdot V}\quad {{s0} \cdot \Delta}\quad {t \cdot \left\lbrack \left\{ {{\left( {{{1/L}\quad d\quad c} + {{1/L}\quad q\quad c}} \right)\cos \quad \theta \quad v} +} \right. \right.}}} \\{{{\quad \left. {\left( {{{1/L}\quad d\quad c} + {{1/L}\quad q\quad c}} \right)\cos \quad \left( {{\theta \quad v} - {2\theta \quad c}} \right)} \right\}}\cos \quad \theta \quad h} +} \\{\quad \left\{ {{\left( {{{1/L}\quad d\quad c} + {{1/L}\quad q\quad c}} \right)\sin \quad \theta \quad v} - \left( {{{1/L}\quad d\quad c} + {{1/L}\quad q\quad c}} \right)} \right.} \\\left. {\left. {\quad \left. {\sin \quad \left( {{\theta \quad v} - {2\theta \quad c}} \right)} \right)} \right\} \sin \quad \theta \quad h} \right\rbrack \\{= \quad {{\left( {1/2} \right) \cdot V}\quad {{s0} \cdot \Delta}\quad {t \cdot \left\lbrack {\left( {{{1/L}\quad d\quad c} + {{1/L}\quad q\quad c}} \right)\cos \quad \theta \quad v} \right.}}} \\{{\quad \left. {{\cos \quad \theta \quad h} + {\sin \quad \theta \quad v\quad \sin \quad \theta \quad h}} \right)} + \left( {{{1/L}\quad d\quad c} - {{1/L}\quad q\quad c}} \right)} \\{\quad \left. \left\{ {{{\cos \left( {{\theta \quad v} - {2\theta \quad c}} \right)}\cos \quad \theta \quad h} - {{\sin \left( {{\theta \quad v} - {2\theta \quad c}} \right)}\sin \quad \theta \quad h}} \right\} \right\rbrack} \\{= \quad {{\left( {1/2} \right) \cdot V}\quad {{s0} \cdot \Delta}\quad {t \cdot \left\lbrack \left( {{{1/L}\quad d\quad c} + {{1/L}\quad q\quad c}} \right) \right.}}} \\{\quad {{\cos \left( {{\theta \quad v} - {\theta \quad h}} \right)} + \left( {{{1/L}\quad d\quad c} - {{1/L}\quad q\quad c}} \right)}} \\{\quad \left. {\cos \left( {{\theta \quad v} - {2\theta \quad c} + {\theta \quad h}} \right)} \right\rbrack} \\{= \quad {{\left( {1/2} \right) \cdot V}\quad {{s0} \cdot \Delta}\quad {t \cdot \left( {{{1/L}\quad d\quad c} + {{1/L}\quad q\quad c}} \right)}}} \\{\quad {\cos \left( {{2\theta \quad v} - {2\theta \quad h} - {\pi/2}} \right)}} \\{= \quad {{\left( {1/2} \right) \cdot V}\quad {{s0} \cdot \Delta}\quad {t \cdot \left( {{{1/L}\quad d\quad c} + {1L\quad q\quad c}} \right)}}} \\{\quad {\sin \quad 2\left( {{\theta \quad v} - {\theta \quad c}} \right)}}\end{matrix}\quad} & {{Formula}\quad 22}\end{matrix}$

[0139] where Vs0 is the absolute value of the voltage differentialvector ΔVs. The absolute value ΔVs0, as shown in FIG. 14(f), varies withthe phase of the control voltage vector, so that the h-axis referencecurrent differential calculation unit 23 obtains the h-axis referencecurrent difference Δihc using the following formula on the basis of thecurrent change per unit voltage.

Δihc=(½)·Δt·(1/Ldc+1/Lqc)sin 2(θv−θc)

[0140] Next, the real current difference Δih in the direction of the haxis is obtained by the h-axis current differential calculation unit 24.

[0141] Step 131

[0142] The V-phase and W-phase current differences Δiva and Δiwa in thesection A are calculated. Symbols iv(ta), iv(tb), and iv(tc) indicateV-phase currents respectively at the times ta, tb, and tc shown in FIG.13. In the same way, symbols iw(ta), iw(tb), and iw(tc) indicate W-phasecurrents respectively at the times ta, tb, and tc.

[0143] Step 132

[0144] The V-phase and W-phase current differences Δivb and Δiwb in thesection B are calculated.

[0145] Step 133

[0146] From the difference between the current differences in thesections A and B, the V-phase and W-phase current differences Δiv andΔiw are obtained. Further, the U-phase current difference Δiu isobtained by calculation of (−Δiv−Δiw). These current differencesindicate changes in the current due to the voltage difference vector ΔVswhich is the difference in the applied voltages in the sections A and B.

[0147] Step 134

[0148] On the basis of the 3-phase current differences Δiu, Δiv, andΔiw, the h-axial phase θh, and the absolute value ΔVs0 of the voltagedifference vector, the h-axial current difference Δih is obtained. Inthis case, division by the absolute value ΔVs0 means the currentdifference per unit voltage.

[0149] The difference between the h-axial current difference Δih and theh-axial reference current difference Δihc obtained in this way is thedifference between the rotor position θ and the inferred rotor positionθc, so that the inferred rotor position θc can be converged to the rotorposition θ using the proportion-integration calculation by the positiondetection unit 13 so as to set the difference between the h-axialcurrent difference Δih and the h-axial reference current difference Δihcto 0. This principle is the same as that of the first embodiment.However, the reason that the function block is complicate is that thecurrent change of the rotatory coordinate system of the h axis isdetected instead of the current change in the direction of the α axis(U-phase direction).

[0150] The detection principle of the rotor position will be explainedin detail by referring to the concrete vector diagram shown in FIG. 17.FIG. 17 shows, in the same way as with FIG. 5, the condition that theinferred rotor position θc inferred by the controller 4 is shifted inthe direction proceeding more than the real rotor position θ. Thevoltage difference vector ΔVs varies with the control voltage vector Vc,so that in FIG. 17, a case that the phase θvc of the control voltagevector Vc is 150 degrees will be explained.

[0151] The voltage difference vector ΔVs in this case, as shown in FIG.14(e), is directed in the direction of −120 degrees. For the voltagedifference vector ΔVs, the real current differential vector Δi and thereference current differential vector Δic are respectively the arrow ofa solid line and the arrow of a dashed line shown in FIG. 17. Therefore,the detection current differential vector Δic, as shown in FIG. 17, isdirected in the direction of the first quadrant. Then, in considerationof the h axis obtained from Formula 21, it is found that it is the thirdquadrant close to −90 degrees. Therefore, the h-axial component of thedetection current differential vector Δis is negative and it may be saidthat the real rotor position θ is smaller than the inferred rotorposition θc. When the value is input to the position detection unit 13,the position detection unit 13 calculates so as to make the inferredrotor position θc smaller, so that it gradually approaches the realrotor position θ.

[0152] Next, even when the control voltage vector Vc is different fromthe example shown in FIG. 17, it will be explained by referring to FIGS.18 and 19 that position detection is possible.

[0153] As shown in FIG. 18(e), when the control voltage vector Vc is at170 degrees, the voltage difference vector ΔVs is directed in thedirection close to −150 degrees. Further, FIG. 18(f) shows that theabsolute value ΔVs0 thereof is made smaller than that shown in FIG. 17.FIG. 19 is a vector diagram at this time.

[0154] The control voltage vector Vc and the voltage difference vectorΔVs are changed compared with the case shown in FIG. 17. Therefore, thecurrent differential vector Δi and the reference current differentialvector Δic are also changed. Actually, the absolute value of the voltagedifference vector ΔVs is reduced and the magnitudes of the currentdifferential vector Δi and the reference current differential vector Δicare also changed. FIGS. 17 and 19 show current changes per unit voltage.This process is the one executed at Step 134 shown in FIG. 16.

[0155] Due to such a relation, the detection current differential vectorΔis is directed in the direction of the second quadrant. On the otherhand, the h axis is in the fourth quadrant, so that the h-axialcomponent of the detection current differential vector Δis has anegative value in the same way as with FIG. 17. Therefore, it is foundthat the inferred rotor position θc at that time proceeds than the realrotor position θ. Namely, when the detection current differencecomponent in the direction of the h axis is detected regardless of thedirection of the control voltage vector Vc, the shift of the rotorposition can be inferred. By this method, the shift of the rotorposition can be detected every period of the PWM signal using thesaliency of the rotor, so that the rotor position can be inferred athigh speed.

[0156] According to the fourth embodiment, from the waveform of the PWMsignal generated by the control voltage, a voltage difference equivalentto the detection voltage can be obtained, so that the current changingcondition for the voltage difference is detected without applying thedetection voltage and the rotor position can be inferred at high speed.Therefore, according to the fourth embodiment, a rapid-response positionsensorless control system can be realized free of noise generated byaddition of the detection voltage and an increase in loss.

[0157] The fifth embodiment of the present invention will be explainedby referring to FIG. 20. The fifth embodiment has a constitution ofchanging the calculation method for detection of the rotor positionaccording to the motor speed ω. The fourth embodiment shown in FIG. 12can detect the rotor position without applying the detection voltage.However, the voltage difference vector equivalent to the detectionvoltage varies with the magnitude of the control voltage, so that in thelow torque operation state at low speed, the voltage difference vectorreduces and the position detection precision lowers. On the other hand,the detection methods shown in FIG. 1 (the first embodiment), FIG. 8(the second embodiment), and FIG. 11 (the third embodiment) are a methodfor applying the detection voltage vector, so that they can detect theposition precisely even in the stop state and low speed state.Therefore, the fifth embodiment has a constitution that the detectionmethod is switched according to the motor speed ω, thus convenientposition detection is realized.

[0158] A main difference of the fifth embodiment from the fourthembodiment shown in FIG. 12 is that a mode decision unit 30 isinstalled, and the calculation contents of the voltage setting unit 25are changed according to the operation mode, and current detection isexecuted by the U-phase current sensor 5 u and the V-phase currentsensor 5 v. Changing of the current sensor is made for simpleexplanation of the fifth embodiment and the phase of current detectionis not limited.

[0159] The function of the mode decision unit 30 will be explained byreferring to the flow chart shown in FIG. 21.

[0160] Step 141

[0161] The mode decision unit 30 inputs the motor speed ω from the speeddetection unit 14.

[0162] Step 142

[0163] The mode decision unit 30 compares the absolute value of themotor speed ω with the first speed ω1 and branches the process.

[0164] Step 143

[0165] When the absolute value of the motor speed ω is not lower thanthe first speed ω1, the mode decision unit 30 compares the absolutevalue with the second speed ω2 and branches the process.

[0166] Step 144

[0167] When the absolute value of the motor speed ω is lower than thefirst speed ω1, the mode decision unit 30 sets the mode MD to 1 (meansthat the synchronous motor 1 is in the low speed state including stop).

[0168] Step 145

[0169] When the absolute value of the motor speed ω is lower than thesecond speed ω2, the mode decision unit 30 sets the mode MD to 2 (meansthat the synchronous motor 1 is in the intermediate speed state).

[0170] Step 146

[0171] When the absolute value of the motor speed ω is the second speedω2 or higher, the mode decision unit 30 sets the mode MD=3 meaning thehigh speed state.

[0172] The PWM signal generation unit 8 and the voltage setting unit 25input the set mode MD.

[0173] The PWM signal generation unit 8, when the mode MD=1, outputs thecarrier synchronizing signal P1 for setting the timing of currentdetection and when the mode MD=2 or MD=3, the PWM signal generation unit8 outputs the carrier synchronizing signal P2. The relation between thecarrier synchronizing signals P1 and P2 and the carrier is as explainedby referring to FIGS. 2 and 13. This means that the mode MD=1 is basedon the method of the first embodiment shown in FIG. 1 and the mode MD=2and MD=3 are based on the method of the fourth embodiment shown in FIG.12.

[0174] The processing function of the voltage setting unit 25 will beexplained by referring to FIG. 22. A difference from the voltage settingunit 25 shown in FIG. 15 is that a plurality of voltage calculationunits and a switching unit are provided. The switching unit 37 selectsand outputs calculation results of the first voltage calculation unit35, the second voltage calculation units 33, and the third voltagecalculation unit 31 according to the mode MD=1, 2, or 3.

[0175] When the speed ω of the synchronous motor 1 is in the low modeMD=1, the voltage setting unit 25 selects the first voltage calculationunit 35 and executes the calculation close to the process of the secondembodiment shown in FIG. 8. The voltage setting unit 25 outputs the sinewave voltages shown in FIG. (a) for the voltage phase θvc, multipliesthem by the absolute value Vc0 of the control voltage vector by themultiplication unit 36, thereby sets the unit voltage of each phaseproportional to the control voltage to the control voltages of eachphase Vuc, Vvc, and Vwc. Further, the voltage setting unit 25 decidesthe detection voltages of each phase Vus, Vvs, and Vws by the inferredrotor position θc by the method shown in FIG. 3 and processes so as tooutput them from the PWM signal generation unit 8 at the timing of thecarrier synchronizing signal P1. The absolute value Vs1 of the detectionvoltage equivalent to the absolute value ΔVs0 of the voltage differencevector is constant, so that this value is output as the absolute valueΔVs0 of the voltage difference vector. Further, the detection voltagedirection θv equivalent to the phase of the voltage difference vector isoutput by calculating (2θc+π/2). As Formula 21 shows, this means thatthe h-axial direction θh is 0, that is, it is the U-phase direction.Therefore, performing of the process of the first voltage calculationunit 35 is performing of the same calculation as that of the secondembodiment shown in FIG. 8.

[0176] When the speed ω of the synchronous motor 1 is in the middlespeed mode MD=2, the voltage setting unit 25 selects the second voltagecalculation unit 33 and executes the calculation. The calculationbasically executes the same process as that of the voltage setting unit25 of the fourth embodiment shown in FIG. 15, though it is a differencethat the table used for calculation is prepared on the basis of FIGS.24(a) and (f).

[0177] In FIG. 24, the voltage of each phase has a waveform that a0-phase voltage of triple harmonic wave is added to the normal sine wavevoltage. The reason of adoption of the waveform is that as shown in FIG.24(f), for the phase θvc of the control voltage vector, the absolutevalue of the voltage difference vector ΔVs is changed little and thephase is changed almost constantly. Therefore, the position detectionprecision can be ensured stably and the control in the middle speedrotation region can be stabilized. Further, the voltages as shown inFIG. 24(a) are ones with the 0-phase voltage added, so that the currentwaveform of each phase will not be adversely affected.

[0178] When the speed ω of the synchronous motor 1 further increases andreaches the high-speed rotation state mode MD=3, the voltage settingunit 25 selects the calculation by the third voltage calculation unit31. In this case, the table for performing the voltage calculation ofeach phase is prepared on the basis of FIGS. 23(a) and (f). As FIG. 23shows, when the phase of the 0-phase voltage of the third harmonic waveis reversed 180 degrees, the maximum value of the voltage of each phaseis reduced. By doing this, when the synchronous motor 1 rotates at highspeed and the counter electromotive force increases, the voltage userate of the inverter 3 can be improved. As shown in FIG. 24(f), there isa disadvantage that the absolute value of the voltage difference vectorvaries greatly, while there is an advantage that the voltage range forrealizing stable control can be enlarged.

[0179] The fifth embodiment can precisely detect the rotor positionwithin the wide range from the stop state to the high-speed rotationstate of the synchronous motor 1 and execute rapid-response control.

[0180] Further, it is basically desirable for this method to togetheruse polarity discrimination for judging the N pole or S pole.

[0181] With respect to the synchronous motor 1 of each embodimentmentioned above, the motor including the rotor having reverse saliencyis explained. However, also to a synchronous motor or a reluctance motorhaving saliency, the present invention can be applied using thesaliency. Further, also to an induction motor, the present invention canbe applied by making the reluctance different between the magnetic fluxdirection and the perpendicular direction to it from the magneticsaturation characteristic by the magnetic flux.

[0182] Further, needless to say, in consideration of the effect byrotation of the rotor of the motor during the sampling time, the poleposition may be calculated.

[0183] Pole position detection is not limited to the method forexecuting every one period or two periods of the carrier and a methodfor detecting the pole position every multi-period of the carrier usingcurrent changes and a method for detecting the pole position on thebasis of current changes in units of a plurality of periods can beexecuted.

[0184] This control not only can be applied to a driving AC motor for anelectric car or a hybrid car but also can be widely applied as aposition sensorless control system of an AC motor.

[0185] The present invention can realize a rapid-response motorcontroller without using a pole position sensor for detecting therotation position of a rotor.

What is claimed is:
 1. A motor control apparatus comprising: an ACmotor, a power conversion device for adding a voltage to said AC motoraccording to a pulse width modulation signal generated by comparing anorder value with a carrier wave, a control device for controlling saidorder value by detecting a position of a rotator of said AC motor basedon a differential between an actual current differential vector and areference current differential vector, and a current sensor abnormalitydetector for outputting a current sensor abnormality signal value byjudging whether a sum of said respective electric current differentialvalues calculated by detecting an electric current from each phase ofsaid AC motor and by detecting an electric current differential value,is more than a predetermined value or not.
 2. A motor control apparatuscomprising: an AC motor, a power conversion device for adding a voltageto said AC motor according to a pulse width modulation signal generatedby comparing an order value with a carrier wave, a current sensorabnormality detector for outputting a current sensor abnormality signalvalue by judging whether a sum of said respective electric currentdifferential values calculated by detecting an electric current fromeach phase of said AC motor and by detecting an electric currentdifferential value, is more than a predetermined value or not, a firstphase current detection part for detecting respective said electriccurrent differential values which change according to each voltagevector added by plural interval, a standard phase current differenceoperation part for operating a standard phase current differenceobtained by difference of plural said voltage vectors, and a positiondetection part for detecting a rotator position of said AC motor usingsaid electric current difference and said standard phase currentdifference, said motor control apparatus characterized by furthercomprising: a control device for controlling said voltage order value bydetecting said rotator position of said AC motor and for controllingsaid AC motor including said current sensor abnormally signal value. 3.A motor control apparatus as defined in claim 2, wherein said firstphase current detection part has a function to remove a DC component. 4.A motor control apparatus as defined in claim 2, comprising: a secondphase current detection part for detecting a phase currents of twophases which is different from that detected by said first phase currentdetection part, wherein said phase current is controlled based on outputof said second phase current detection part.
 5. A motor controlapparatus as defined in claim 4, said control device comprisesabnormality detection part for detecting an abnormality of said first orsecond phase current detection part based on an alternating component ofsaid phased currents detected by said first and second phase currentdetection parts.